The present invention relates to a power conversion circuit and more particularly to a power conversion circuit that is capable of more effectively making use of the high-speed characteristics of a switching device compared with a related power conversion circuit using a switching device capable of high-speed switching with the same breakdown voltage and rated current as those of the switching device used in the power conversion circuit according to the invention.
A high breakdown voltage power device fabricated by using a silicon carbide semiconductor (hereinafter also referred to as SiC) with a band gap wider than that of silicon (hereinafter referred to as Si) or a group III nitride semiconductor (hereinafter also referred to as GaN, AlGaN, etc.) has a possibility of significantly decreasing on-state resistance. With a MISFET (Metal-Insulator-Semiconductor Field Effect Transistor) of breakdown voltage of 1 kV to 1.2 kV class, on-state resistance of 5 mΩcm2 or less is obtained. The on-state resistance is equal to a half or less compared with the on-state resistance of an IGBT (Insulated Gate Bipolar Transistor) of Si of the equal class of breakdown voltage. There is therefore the possibility that future progress in cost reduction and characteristics improvement will lead to substitution of power devices using SiC for most of IGBTs of silicon as a component of an inverter.
The reason for enabling significant decrease in on-state resistance by using SiC, AlGaN, etc. is that the dielectric breakdown electric field strength of SiC, AlGaN, etc. is so high that a voltage withstanding layer can be made thin and the amount of doping can be increased to thereby enable the resistance of the voltage withstanding layer to be decreased by two digits or more compared with the resistance of the voltage withstanding layer of Si.
For forming a circuit using such high break down voltage power devices, studies have been carried out using a circuit formed using IGBTs or a circuit formed by using Si MOSFETs (Metal-Oxide-Semiconductor Field Effect Transistors) for small capacities and (relatively) low voltages.
FIG. 10 is a circuit diagram showing a power conversion circuit for a related hard switching system, FIG. 11 is a circuit diagram showing an example of the circuit of a related multi-level converter, FIG. 12 is a circuit diagram showing an example of the circuit for a related voltage resonance type soft switching system, and FIG. 13 is a circuit diagram showing an example of the circuit for a related current resonance type soft switching system.
The power conversion circuit shown in FIG. 10 is a circuit for a so-called hard switching system which previously has been widely fabricated using IGBTs. The power conversion circuit has paired switching devices M1 and M2 provided between the positive and negative poles of a DC power supply V0. In parallel to the switching device M1, a free wheeling diode D1 is provided and, in parallel to the switching device M2, a free wheeling diode D2 is provided. When the switching devices are IGBTs, no current flows in the reverse direction (upward in FIG. 10), which requires external free wheeling diodes D1 and D2. However, when the switching devices are MOSFETs, internally formed body diodes can be used as free wheeling diodes D1 and D2. To an output point NM as the mid-point of the switching devices M1 and M2, a first end of an inductive load LL is connected. For simplicity, FIG. 10 depicts only one phase of the circuit, so that the output end (second end) of the load LL seems to be connected nowhere. Actually, in a single phase system, the second end of the load LL is to be connected to the output point of a power conversion circuit having the same configuration as that shown in FIG. 10; and, in a multiple-phase system with three or more phases, the second ends of similar loads, connected to their respective power conversion circuits each having the same configuration as that shown in FIG. 10, are to be connected together into one.
In general, semiconductor devices provide better properties when using either conduction type of semiconductor material in a particular portion than when using the other type therein. Hence, devices using the same conduction type therein are used for the switching devices M1 and M2. For example, of MOSFETS using Si as the semiconductor material, n-type devices are usually employed, wherein n-type is assigned for the voltage withstanding layer and inversion channel regions owing to the higher mobility of majority carriers in n-type.
In the circuit shown in FIG. 10, n-type devices are used for the switching devices M1 and M2 in each of which the drain is connected to the positive potential side to the potential of the source. Thus, as shown in FIG. 10, the reference potential of a control circuit G2, that controls the switching device M2 connected to the section between the negative pole side of the DC power supply V0 and the output point NM (referred to as a lower arm), becomes the potential of the negative pole side of the DC power supply V0. Compared with this, the reference potential of a control circuit G1, that controls the switching device M1 connected to the section between the positive pole side of the DC power supply V0 and the output point NM (referred to as an upper arm), becomes the potential of the output point NM at a potential on the positive side to the reference potential of the control circuit G2. For controlling the switching speeds of the switching devices M1 and M2, between the switching device M1 and the control circuit G1 and between the switching device M2 and the control circuit G2, gate resistors R1 and R2 are provided, respectively. The resistance value of a gate resistor at turning-on at which a switching device is brought from a conductive state to a blocking state is generally different from the value at turning-off at which the switching device is brought from the blocking state to the conductive state. However, for simplicity, the configuration for this purpose is omitted in FIG. 10.
The potential at the output point NM varies from the potential of the negative pole of the DC power supply V0 to the potential of its positive pole. This necessitates control signals given to the control circuits G1 and G2 to be electrically insulated. Thus, in a high voltage power conversion circuit, a signal transmitting device with an insulating property such as a photocoupler is used. Such a device, however, has a possibility of causing a malfunction at an abrupt change in its reference potential. Therefore, in general, a certain limitation is imposed in that the rate of change of the potential at the output point NM with respect to time (hereinafter the rate of change of a potential or a voltage with respect to time will be sometimes abbreviated as dV/dt regardless of the potential or the voltage) is to be, for example, 10 kV/μs or less.
In a power conversion circuit, there is also a limitation on the rate of change of a current (hereinafter also referred to as dI/dt) flowing in a switching device. Although some kinds of switching devices (IGBTs, etc.) are broken down by excessively high dI/dt that causes local concentration of currents, the limitations on dI/dt are also imposed on circuit configurations. One of them is a limitation on dI/dt due to a voltage surge produced at turning-off. For example, in the case of turning-off the switching device M2, in a closed circuit including the DC power supply V0, the load LL (at the turning-off of the switching device M2, the second end of the load LL is equivalently connected to the positive pole of the DC power supply V0) and the switching device M2, there exists stray inductance not shown in FIG. 10. Turning-off of the switching device M2 causes a current flowing in the load LL to flow in the free wheeling diodes D1 and thus cause no particular problem. However, a current flowing through the stray inductance decreases with time to produce an induced electromotive force proportional to the value of the stray inductance and the value of dI/dt. Thus, the voltage due to the electromotive force is superimposed on the voltage at the output point NM to be a voltage surge. A voltage surge causes insulation breakdown on the load side, at worst. Therefore, there is a certain limitation on the value of the voltage surge. From the viewpoint of this, in an ordinary power conversion circuit, care is taken so that the least possible inductance is present.
Incidentally, the so-called multi-level converter exemplified in FIG. 11 might be regarded as an exception. The multi-level converter is used for a purpose of treating a very high power supply voltage. In this case, a number of switching units SUs are paired on the positive pole side and the negative pole side of the DC power supply V0 while being connected in series. Between the series connected switching units SUs on the positive pole side of the DC power supply V0 and the output point NM, an inductor L1 is provided, and between the series connected switching units SUs on the negative pole side of the DC power supply V0 and the output point NM, an inductor L2 is provided. The reason why the multi-level converter unexpectedly has the inductors L1 and L2 connected to the output point NM is that the requirement for each of the switching units is to have a property of allowing a current to flow with an arbitrary amount within the rated amount regardless of a voltage across the unit. No single device is known that meets the requirement and, in many cases, the switching unit SU itself is further formed of a different power conversion circuit. For example, the switching unit SU can include the power conversion circuit according to the invention. Accordingly, a multi-level converter is to be regarded as having an objective different from that of the invention.
Furthermore, a power conversion device is also known which is formed by connecting a switching device and a reactor in a DC current path between the positive pole and the negative pole of a DC power supply (see JP-A-2009-11117 , for example). To the reactor, a free wheeling diode is connected in parallel. Also in the power conversion circuit, however, the reactor connected to the switching device is provided for the purpose of detecting an abnormality due to a short circuit between the positive pole and the negative pole of the DC power supply. Thus, a circuit configuration is provided in which a flow of an overcurrent in the reactor due to an abnormality such as a short circuit of the switching device causes a reverse voltage to be applied to the free wheeling diode and the reverse voltage is detected by a controller to be determined as an occurrence of an abnormality to turn-off the switching device. Therefore, the power conversion device disclosed in JP-A-2009-11117 uses the reactor for detecting an overcurrent state of the main current, the objective of which is thus entirely different from that of the invention.
As was described in the foregoing, there are limitations in values of dV/dt and dI/dt. Thus, there is generally an upper limit to the operation speed of a switching device. However, a switching speed that is too low generally prolongs the period within which a current flows in the switching device with a non-zero voltage kept applied to the switching device, due to which a switching loss unfavorably increases. For adjusting the switching speed, adjustment of gate resistance is generally carried out.
Here, an explanation will be made with respect to gate resistance adjustment in the case in which the switching device M2 shown in FIG. 10 is a trench MOSFET of SiC taken as an example for simplicity.
At the turning-off of the switching device M2, the decrease in the output voltage of the control circuit G2 causes the source-gate capacitor of the switching device M2 to discharge through the gate resistor R2, which decreases a source-gate voltage (hereinafter simply referred to as a gate voltage). Then, a current flowing between the source and the drain (hereinafter simply referred to as a drain current) of the switching device M2 is to decrease. The value of the current must be equal to the value of the current flowing in the inductive load LL. Thus, for inhibiting the decrease in current, an induced electromotive force is produced in the load LL and a voltage due to the induced electromotive force is applied as the source-drain voltage (hereinafter simply referred to as the drain voltage) of the switching device M2. A change in the time of the drain voltage (dV/dt) of the switching device M2 causes a displacement current to flow through a gate-drain capacitor with a value proportional to both of the gate-drain capacitance and the change with respect to time in the drain voltage. Since the displacement current causes a voltage drop across the gate resistor R2, no drop occurs in the gate voltage of the switching device M2. In the case of the circuit on the hard switching system shown in FIG. 10, the gate voltage of the switching device is maintained so that at least the current flowing before the turning-off can be maintained. For maintaining the gate voltage, the value of dV/dt is determined. The larger the value of the gate resistor R is or the larger the gate-drain capacitance of the switching device M2 is, the smaller the value of dV/dt for maintaining the same voltage becomes. When the value of the potential at the output point NM becomes higher than the value of the potential on the positive pole side of the DC power supply V0 (more precisely, the value with the on-state voltage of the free wheeling diode added to the value of the potential on the positive pole side), the free wheeling diode D1 is rendered conductive. Thus, the potential at the output point NM is fixed at the potential on the positive pole side of the DC power supply to make the value of dV/dt zero. This causes the gate voltage of the switching device M2 to no longer be maintained in a manner to decrease the gate voltage with a time constant determined by the source-gate capacitance and the resistance of the gate resistor R2. Until the gate voltage decreases to a threshold voltage or less, a drain current continues to flow with its value corresponding to the gate voltage. The larger the resistance of the gate resistor R2 becomes, the less sharp the fall characteristic of the drain current generally becomes.
At the turning-on of the switching device M2, the turning-on starts from a state in which a specified current flows in the load LL. An increase in the gate voltage of the switching device M2 with a time constant determined by the source-gate capacitance and the resistance of the gate resistor R2 increases the drain current according to the increase in the gate voltage. However, until the value of the drain current reaches the specified value of the current flowing in the load LL, a current with the value being equivalent to the difference from the specified value must flow through the free wheeling diode D1. Hence, the potential at the output point NM is kept at the potential at turning-off except the potential due to the induced electromotive force produced with an increase in the drain current by the presence of stray inductance. When the value of the drain current in the switching device M2 reaches the specified current value, the free wheeling diode D1 (for which a diode with significantly small reverse recovery such as a Schottky barrier diode is to be used) turns off to decrease the drain voltage of the switching device M2. Like at the turning-on, the value of dV/dt is determined so that no gate voltage falls below the gate voltage necessary for maintaining the drain current.
As was explained in the foregoing, by increasing the value of the resistance of the gate resistor R2 according to the gate-drain capacitance of the switching device M2, the value of dV/dt can be prevented from becoming too large. However, the value of the resistance of the gate resistor R2 affects the sharpness in falling-off a current after the drain voltage reaches the value at which the switching device M2 is in a turn-off state (the value as a sum of the voltage of the DC power supply and the on-state voltage of the free wheeling diode D1) and the rising of a current before the drain voltage reaches the value at which the switching device M2 is in a blocking state. Therefore, the value of the resistance is excessively high and causes a current to fall off with poor sharpness and slow rise, which result in an increase in a switching loss.
Incidentally, as a circuit that can decrease a turn-off loss without unintentionally increasing the value of dV/dt, there is known a power conversion circuit on a voltage resonance type soft switching system such as that shown in FIG. 12. The difference from the hard switching system is that capacitors C1 and C2 are provided in parallel to the switching devices M1 and M2, respectively (when the circuit shown in FIG. 12 is independently used, only one of the capacitors C1 and C2 can be used).
For example, when the switching device M2 is turned-off, unlike a circuit on a hard switching system, it is necessary only that the sum of the value of the drain current of the switching device M2 and the value of a current charging the capacitor C2 are kept equal to the value of the current flowing the load LL. Therefore, when the capacitance of the capacitor C2 is sufficiently large, before the drain voltage of the switching device M2 (equal to the voltage across the capacitor C2) begins to significantly increase, the gate voltage decrease to the threshold voltage or below, by which the drain current becomes zero. At this time, a turn-on loss virtually becomes zero and, along with this, the value of dV/dt becomes the value of a quotient for which the value of the current flowing in the load LL is divided by the value of the capacitance of the capacitor C2 and no dV/dt is produced with a value more than the quotient. Even though the situation is not extreme to such an extent, the drain current decreases by an amount of a current charging the capacitor C2 to decrease the turn-off loss. Here, since no value of dV/dt exceeds its allowed value, the turn-off loss is to be decreased without unintentionally increasing the value of dV/dt.
However, in the circuit as is shown in FIG. 12, all of currents due to the discharge of the capacitor C2 (and the capacitor C1) at turning-on flow in one switching device (M2 or M1) to increase a current load imposed on the switching device. Along with this, a switching (turn-on) loss also increases. In particular, the turn-on loss is generally larger than the sum of a turn-on loss when a circuit on a hard switching system is turned-on with the same gate resistor R2 used and energy stored in the capacitor C2 (and the capacitor C1). This is because the value of dV/dt is too small for maintaining a current due to discharge of the capacitor C2 (and the capacitor C1).
Conversely, as a circuit for decreasing a turn-on loss, there is known a power converting circuit on a current resonance type soft switching system. The power conversion circuit, with various problems being ignored, is presented as, in the simplest way, the circuit as shown in FIG. 13, for example. The difference with respect to the circuit on the hard switching system shown in FIG. 10 is that an inductor L0 is provided between the DC power supply V0 and the power conversion circuit on the hard switching system. The inductor L0 does not necessarily need to be provided between the DC power supply V0 and the switching device M1. For example, for decreasing the turn-on loss in the switching device M2, the inductor L0 can be provided somewhere in a closed circuit including no load LL and no free wheeling diode D1. At turning-on, a sufficiently large inductance of the inductor L0 allows the maximum value of an induced electromotive force produced in the inductor L0 to be equal to the voltage of the DC power supply V0 (with other small voltages being ignored), by which the drain voltage of the switching device M2 becomes approximately zero. Thereafter, the current in the switching device M2 increases with a current increasing rate that the voltage of the DC power supply V0 is divided by the inductance of the inductor L0. At this time, a turn-on loss virtually becomes zero. Even though the state is not extreme to such an extent, the voltage due to the induced electromotive force produced in the inductor L0 shares the voltage of the DC power supply V0 together with the drain voltage of the switching device M2. Thus, the drain voltage of the switching device M2 is decreased by the voltage produced in the inductor L0 to decrease the turn-on loss by the amount corresponding to the decrease in the drain voltage.
The circuit as shown in FIG. 13 is equivalent to a hard switching circuit with significantly increased stray inductance. Thus, production of a significantly large surge voltage is unavoidable at turning-off.
Here, for reference, explanations will be made with respect to specific examples of switching losses in the above power conversion circuit on the hard switching system (the power conversion circuit shown in FIG. 10) and the power conversion circuit on the soft switching system (represented by the power conversion circuit shown in FIG. 12).
FIG. 14 is a diagram showing an example of the turn-off loss in a power conversion circuit on a hard switching system, FIG. 15 is a diagram showing an example of the turn-on loss in the power conversion circuit on the hard switching system, FIG. 16 is a diagram showing an example of the turn-off loss in a power conversion circuit on a soft switching system and FIG. 17 is a diagram showing an example of the turn-on loss in the power conversion circuit on the soft switching system.
Here, as an example of each of the switching devices M1 and M2, a SiC trench MOSFET with a rated gate voltage of 42V and a rated current of 75 A is used which is arranged to carry out switching of a DC power supply V0 with a power supply voltage of 600V. Each of FIG. 14 and FIG. 16 shows plots with respect to a gate resistance of a turn-off loss (P_off) represented by a broken line and the maximum dV/dt (the maximum value of dV/dt at switching) represented by a solid line. Each of FIG. 15 and FIG. 17 shows plots with respect to a gate resistance of a turn-on loss (P_on) represented by a broken line and the maximum dV/dt represented by a solid line.
First, with reference to FIG. 14 and FIG. 15, an explanation will be made with respect to the switching loss in power conversion circuits on a hard switching system. The value of dV/dt decreases as the gate resistor is made larger. However, at turning-off, even in the case of letting the gate resistance be 50Ω, it is still impossible to reduce the maximum dV/dt to its allowed value of, for example, 10 kV/μs or less. On the other hand, increased gate resistance increases both of turn-off loss and turn-on loss. For example, when gate resistance is 50Ω, a turn-off loss becomes approximately 2 mJ/pulse and a turn-on loss becomes approximately 4 mJ/pulse, and the sum of them becomes approximately 6 mJ/pulse.
Ordinarily, in the case of using an IGBT with the above rated voltage and current, with the use of a Schottky barrier diode for the free wheeling diode, even with the sum of the turn-off loss and the turn-on loss, the switching loss is 10 mJ/pulse or less to be, for example, on the order of 5 mJ/pulse, though the losses depend on various circuit conditions. A MOSFET has the advantage of being faster in switching speed compared with an IGBT. However, in the power conversion circuit on the hard switching system such as that shown in FIG. 10, the use of a MOSFET is to result in an increase in a switching loss in reverse.
In the following, an explanation will be made concerning the switching loss in a power conversion circuit on a soft switching system. With respect to the case in which a capacitor is provided in parallel to a MOSFET like the power conversion circuit shown in FIG. 12 with the capacitance of the capacitor taken as, for example, 7.5 nF, the results of plots similar to those in the above are shown in FIG. 16 and FIG. 17. From FIG. 16, it is known that, in the case of turning-off, with gate resistance taken as 50Ω, the maximum dV/dt can be decreased to 10 kV/μs or less with the turn-off loss at this time significantly smaller than 1 mJ/pulse. The reason why the maximum dV/dt is high when the gate resistance is low is considered to be an influence of stray inductance in the wiring of the MOSFET (the wiring on the MOSFET side rather than the wiring at the branch of the capacitor C1 or C2).
However, as is understood from FIG. 17, in the case of turning-on, the gate resistance is required to be, for example, 50Ω for decreasing the maximum dV/dt to, for example, 10 kV/μs or less. This results in the switching loss largely exceeding 5 mJ/pulse. With the gate resistance made to be on the order of 30Ω, for example, the turn-on loss might become on the order of 3 to 4 mJ/pulse on the diagram. Further detailed studies, however, show that the turn-on loss sometimes becomes a little lower due to the resonance caused by the presence of the capacitors C1 and C2 and stray inductance (when the gate resistance is 5Ω, for example, the turn-on loss is on the order of 1.2 mJ/pulse, which shows that the energy to be released from the capacitor C2 by turn-on is smaller than 1.35 mJ/pulse). Thus, the turn-on loss is not always become on the order of 3 to 4 mJ/pulse.
That is, when a MOSFET is used for a switching device, even with the use of a power conversion circuit on a voltage resonance type soft switching system, a decrease in switching loss is almost impossible.
[Patent Document 1] JP-A-2009-11117
As was explained in the foregoing, there were problems in that a power conversion circuit on a hard switching system can not decrease the value of the rate of change in voltage to a specified value or less even though gate resistance is adjusted and a power conversion circuit on a voltage resonance type soft switching system can not decrease a switching loss at turning-on of a switching device.
The invention was made in view of such points with an object of providing a power conversion circuit which can decrease the value of a rate of change in voltage without significantly increasing a switching loss when using a switching device that cannot effectively decrease the value of a rate of change in voltage by normally increasing gate resistance to cause the rate of change in voltage to become excessively high.
Further object and advantages of the invention will be apparent from the following description of the invention.